1. Field
The present disclosure relates generally to data networking and in particular to a backhaul radio for connecting remote edge access networks to core networks.
2. Related Art
Data networking traffic has grown at approximately 100% per year for over 20 years and continues to grow at this pace. Only transport over optical fiber has shown the ability to keep pace with this ever-increasing data networking demand for core data networks. While deployment of optical fiber to an edge of the core data network would be advantageous from a network performance perspective, it is often impractical to connect all high bandwidth data networking points with optical fiber at all times. Instead, connections to remote edge access networks from core networks are often achieved with wireless radio, wireless infrared, and/or copper wireline technologies.
Radio, especially in the form of cellular or wireless local area network (WLAN) technologies, is particularly advantageous for supporting mobility of data networking devices. However, cellular base stations or WLAN access points inevitably become very high data bandwidth demand points that require continuous connectivity to an optical fiber core network.
When data aggregation points, such as cellular base station sites, WLAN access points, or other local area network (LAN) gateways, cannot be directly connected to a core optical fiber network, then an alternative connection, using, for example, wireless radio or copper wireline technologies, must be used. Such connections are commonly referred to as “backhaul.”
Many cellular base stations deployed to date have used copper wireline backhaul technologies such as T1, E1, DSL, etc. when optical fiber is not available at a given site. However, the recent generations of HSPA+ and LTE cellular base stations have backhaul requirements of 100 Mb/s or more, especially when multiple sectors and/or multiple mobile network operators per cell site are considered. WLAN access points commonly have similar data backhaul requirements. These backhaul requirements cannot be practically satisfied at ranges of 300 m or more by existing copper wireline technologies. Even if LAN technologies such as Ethernet over multiple dedicated twisted pair wiring or hybrid fiber/coax technologies such as cable modems are considered, it is impractical to backhaul at such data rates at these ranges (or at least without adding intermediate repeater equipment). Moreover, to the extent that such special wiring (i.e., CAT 5/6 or coax) is not presently available at a remote edge access network location; a new high capacity optical fiber is advantageously installed instead of a new copper connection.
Rather than incur the large initial expense and time delay associated with bringing optical fiber to every new location, it has been common to backhaul cell sites, WLAN hotspots, or LAN gateways from offices, campuses, etc. using microwave radios. An exemplary backhaul connection using the microwave radios 132 is shown in FIG. 1. Traditionally, such microwave radios 132 for backhaul have been mounted on high towers 112 (or high rooftops of multi-story buildings) as shown in FIG. 1, such that each microwave radio 132 has an unobstructed line of sight (LOS) 136 to the other. These microwave radios 132 can have data rates of 100 Mb/s or higher at unobstructed LOS ranges of 300 m or longer with latencies of 5 ms or less (to minimize overall network latency).
Traditional microwave backhaul radios 132 operate in a Point-to-point (PTP) configuration using a single “high gain” (typically >30 dBi or even >40 dBi) antenna at each end of the link 136, such as, for example, antennas constructed using a parabolic dish. Such high gain antennas mitigate the effects of unwanted multipath self-interference or unwanted co-channel interference from other radio systems such that high data rates, long range and low latency can be achieved. These high gain antennas however have narrow radiation patterns.
Furthermore, high gain antennas in traditional microwave backhaul radios 132 require very precise, and usually manual, physical alignment of their narrow radiation patterns in order to achieve such high performance results. Such alignment is almost impossible to maintain over extended periods of time unless the two radios have a clear unobstructed line of sight (LOS) between them over the entire range of separation. Furthermore, such precise alignment makes it impractical for any one such microwave backhaul radio to communicate effectively with multiple other radios simultaneously (i.e., a “point-to-multipoint” (PMP) configuration).
In wireless edge access applications, such as cellular or WLAN, advanced protocols, modulation, encoding and spatial processing across multiple radio antennas have enabled increased data rates and ranges for numerous simultaneous users compared to analogous systems deployed 5 or 10 years ago for obstructed LOS propagation environments where multipath and co-channel interference were present. In such systems, “low gain” (usually <6 dBi) antennas are generally used at one or both ends of the radio link both to advantageously exploit multipath signals in the obstructed LOS environment and allow operation in different physical orientations as would be encountered with mobile devices. Although impressive performance results have been achieved for edge access, such results are generally inadequate for emerging backhaul requirements of data rates of 100 Mb/s or higher, ranges of 300 m or longer in obstructed LOS conditions, and latencies of 5 ms or less.
In particular, “street level” deployment of cellular base stations, WLAN access points or LAN gateways (e.g., deployment at street lamps, traffic lights, sides or rooftops of single or low-multiple story buildings) suffers from problems because there are significant obstructions for LOS in urban environments (e.g., tall buildings, or any environments where tall trees or uneven topography are present).
FIG. 1 illustrates edge access using conventional unobstructed LOS PTP microwave radios 132. The scenario depicted in FIG. 1 is common for many 2nd Generation (2G) and 3rd Generation (3G) cellular network deployments using “macrocells”. In FIG. 1, a Cellular Base Transceiver Station (BTS) 104 is shown housed within a small building 108 adjacent to a large tower 112. The cellular antennas 116 that communicate with various cellular subscriber devices 120 are mounted on the towers 112. The PTP microwave radios 132 are mounted on the towers 112 and are connected to the BTSs 104 via an nT1 interface. As shown in FIG. 1 by line 136, the radios 132 require unobstructed LOS.
The BTS on the right 104a has either an nT1 copper interface or an optical fiber interface 124 to connect the BTS 104a to the Base Station Controller (BSC) 128. The BSC 128 either is part of or communicates with the core network of the cellular network operator. The BTS on the left 104b is identical to the BTS on the right 104a in FIG. 1 except that the BTS on the left 104b has no local wireline nT1 (or optical fiber equivalent) so the nT1 interface is instead connected to a conventional PTP microwave radio 132 with unobstructed LOS to the tower on the right 112a. The nT1 interfaces for both BTSs 104a, 104b can then be backhauled to the BSC 128 as shown in FIG. 1.
FIG. 2A is a block diagram of the major subsystems of a conventional PTP microwave radio 200A for the case of Time-Division Duplex (TDD) operation, and FIG. 2B is a block diagram of the major subsystems of a conventional PTP microwave radio 200B for the case of Frequency-Division Duplex (FDD) operation.
As shown in FIG. 2A and FIG. 2B, the conventional PTP microwave radio traditionally uses one or more (i.e. up to “n”) T1 interfaces 204A and 204B (or in Europe, E1 interfaces). These interfaces (204A and 204B) are common in remote access systems such as 2G cellular base stations or enterprise voice and/or data switches or edge routers. The T1 interfaces are typically multiplexed and buffered in a bridge (e.g., the Interface Bridge 208A, 208B) that interfaces with a Media Access Controller (MAC) 212A, 212B.
The MAC 212A, 212B is generally denoted as such in reference to a sub-layer of Layer 2 within the Open Systems Interconnect (OSI) reference model. Major functions performed by the MAC include the framing, scheduling, prioritizing (or “classifying”), encrypting and error checking of data sent from one such radio at FIG. 2A or FIG. 2B to another such radio. The data sent from one radio to another is generally in a “user plane” if it originates at the T1 interface(s) or in the “control plane” if it originates internally such as from the Radio Link Controller (RLC) 248A, 248B shown in FIG. 2A or FIG. 2B.
With reference to FIGS. 2A and 2B, the Modem 216A, 216B typically resides within the “baseband” portion of the Physical (PHY) layer 1 of the OSI reference model. In conventional PTP radios, the baseband PHY, depicted by Modem 216A, 216B, typically implements scrambling, forward error correction encoding, and modulation mapping for a single RF carrier in the transmit path. In receive, the modem typically performs the inverse operations of demodulation mapping, decoding and descrambling. The modulation mapping is conventionally Quadrature Amplitude Modulation (QAM) implemented with In-phase (I) and Quadrature-phase (Q) branches.
The Radio Frequency (RF) 220A, 220B also resides within the PHY layer of the radio. In conventional PTP radios, the RF 220A, 220B typically includes a single transmit chain (Tx) 224A, 224B that includes I and Q digital to analog converters (DACs), a vector modulator, optional upconverters, a programmable gain amplifier, one or more channel filters, and one or more combinations of a local oscillator (LO) and a frequency synthesizer. Similarly, the RF 220A, 220B also typically includes a single receive chain (Rx) 228A, 228B that includes I and Q analog to digital converters (ADCs), one or more combinations of an LO and a frequency synthesizer, one or more channel filters, optional downconverters, a vector demodulator and an automatic gain control (AGC) amplifier. Note that in many cases some of the one or more LO and frequency synthesizer combinations can be shared between the Tx and Rx chains.
As shown in FIGS. 2A and 2B, conventional PTP radios 200A, 200B also include a single power amplifier (PA) 232A, 232B. The PA 232A, 232B boosts the transmit signal to a level appropriate for radiation from the antenna in keeping with relevant regulatory restrictions and instantaneous link conditions. Similarly, such conventional PTP radios 232A, 232B typically also include a single low-noise amplifier (LNA) 236, 336 as shown in FIGS. 2A and 2B. The LNA 236A, 236B boosts the received signal at the antenna while minimizing the effects of noise generated within the entire signal path.
As described above, FIG. 2A illustrates a conventional PTP radio 200A for the case of TDD operation. As shown in FIG. 2A, conventional PTP radios 200A typically connect the antenna 240A to the PA 232A and LNA 236A via a band-select filter 244A and a single-pole, single-throw (SPST) switch 242A.
As described above, FIG. 2B illustrates a conventional PTP radio 200B for the case of FDD operation. As shown in FIG. 2B, in conventional PTP radios 200B, then antenna 240B is typically connected to the PA 232B and LNA 236B via a duplexer filter 244B. The duplexer filter 244B is essentially two band-select filters (tuned respectively to the Tx and Rx bands) connected at a common point.
In the conventional PTP radios shown in FIGS. 2A and 2B, the antenna 240A, 240B is typically of very high gain such as can be achieved by a parabolic dish so that gains of typically >30 dBi (or even sometimes >40 dBi), can be realized. Such an antenna usually has a narrow radiation pattern in both the elevation and azimuth directions. The use of such a highly directive antenna in a conventional PTP radio link with unobstructed LOS propagation conditions ensures that the modem 216A, 216B has insignificant impairments at the receiver (antenna 240A, 240B) due to multipath self-interference and further substantially reduces the likelihood of unwanted co-channel interference due to other nearby radio links.
Although not explicitly shown in FIGS. 2A and 2B, the conventional PTP radio may use a single antenna structure with dual antenna feeds arranged such that the two electromagnetic radiation patterns emanated by such an antenna are nominally orthogonal to each other. An example of this arrangement is a parabolic dish. Such an arrangement is usually called dual-polarized and can be achieved either by orthogonal vertical and horizontal polarizations or orthogonal left-hand circular and right-hand circular polarizations.
When duplicate modem blocks, RF blocks, and PA/LNA/switch blocks are provided in a conventional PTP radio, then connecting each PHY chain to a respective polarization feed of the antenna allows theoretically up to twice the total amount of information to be communicated within a given channel bandwidth to the extent that cross-polarization self-interference can be minimized or cancelled sufficiently. Such a system is said to employ “dual-polarization” signaling. Such systems may be referred to as having two “streams” of information, whereas multiple input multiple output (MIMO) systems utilizing spatial multiplexing may achieve successful communications using even more than two streams, in practice.
When an additional circuit (not shown) is added to FIG. 2A that can provide either the RF Tx signal or its anti-phase equivalent to either one or both of the two polarization feeds of such an antenna, then “cross-polarization” signaling can be used to effectively expand the constellation of the modem within any given symbol rate or channel bandwidth. With two polarizations and the choice of RF signal or its anti-phase, then an additional two information bits per symbol can be communicated across the link. Theoretically, this can be extended and expanded to additional phases, representing additional information bits. At the receiver, for example, a circuit (not shown) could detect if the two received polarizations are anti-phase with respect to each other, or not, and then combine appropriately such that the demodulator in the modem block can determine the absolute phase and hence deduce the values of the two additional information bits. Cross-polarization signaling has the advantage over dual-polarization signaling in that it is generally less sensitive to cross-polarization self-interference but for high order constellations such as 64-QAM or 256-QAM, the relative increase in channel efficiency is smaller.
In the conventional PTP radios shown in FIGS. 2A and 2B, substantially all the components are in use at all times when the radio link is operative. However, many of these components have programmable parameters that can be controlled dynamically during link operation to optimize throughput and reliability for a given set of potentially changing operating conditions. The conventional PTP radios of FIGS. 2A and 2B control these link parameters via a Radio Link Controller (RLC) 248A, 248B. The RLC functionality is also often described as a Link Adaptation Layer that is typically implemented as a software routine executed on a microcontroller within the radio that can access the MAC 212A, 212B, Modem 216A, 216B, RF 220A, 220B and/or possibly other components with controllable parameters. The RLC 248A, 248B typically can both vary parameters locally within its radio and communicate with a peer RLC at the other end of the conventional PTP radio link via “control frames” sent by the MAC 212A, 212B with an appropriate identifying field within a MAC Header.
Typical parameters controllable by the RLC 248A, 248B for the Modem 216A, 216B of a conventional PTP radio include encoder type, encoding rate, constellation selection and reference symbol scheduling and proportion of any given PHY Protocol Data Unit (PPDU). Typical parameters controllable by the RLC 248A, 248B for the RF 220A, 220B of a conventional PTP radio include channel frequency, channel bandwidth, and output power level. To the extent that a conventional PTP radio employs two polarization feeds within its single antenna, additional parameters may also be controlled by the RLC 248A, 248B as self-evident from the description above.
In conventional PTP radios, the RLC 248A, 248B decides, usually autonomously, to attempt such parameter changes for the link in response to changing propagation environment characteristics such as, for example, humidity, rain, snow, or co-channel interference. There are several well-known methods for determining that changes in the propagation environment have occurred such as monitoring the receive signal strength indicator (RSSI), the number of or relative rate of FCS failures at the MAC 212A, 212B, and/or the relative value of certain decoder accuracy metrics. When the RLC 248A, 248B determines that parameter changes should be attempted, it is necessary in most cases that any changes at the transmitter end of the link become known to the receiver end of the link in advance of any such changes. For conventional PTP radios, and similarly for many other radios, there are at least two well-known techniques which in practice may not be mutually exclusive. First, the RLC 248A, 248B may direct the PHY, usually in the Modem 216A, 216B relative to FIGS. 2A and 2B, to pre-pend a PHY layer convergence protocol (PLCP) header to a given PPDU that includes one or more (or a fragment thereof) given MPDUs wherein such PLCP header has information fields that notify the receiving end of the link of parameters used at the transmitting end of the link. Second, the RLC 248A, 248B may direct the MAC 212A, 212B to send a control frame, usually to a peer RLC 248A, 248B, including various information fields that denote the link adaptation parameters either to be deployed or to be requested or considered.
The foregoing describes at an overview level the typical structural and operational features of conventional PTP radios which have been deployed in real-world conditions for many radio links where unobstructed (or substantially unobstructed) LOS propagation was possible. The conventional PTP radio on a whole is completely unsuitable for obstructed LOS PTP or PMP operation.
More recently, as briefly mentioned, there has been significant adoption of so-called multiple input multiple output (MIMO) techniques, which utilize spatial multiplexing of multiple information streams between a plurality of transmission antennas to a plurality of receive antennas. The adoption of MIMO has been most beneficial in wireless communication systems for use in environments having significant multipath scattering propagation. One such system is IEEE802.11n for use in home networking. Attempts have been made to utilize MIMO and spatial multiplexing in line of sight environments having minimal scattering, which have generally been met with failure, in contrast to the use of cross polarized communications. For example IEEE802.11n based Mesh networked nodes deployed at streetlight elevation in outdoor environments often experience very little benefit from the use of spatial multiplexing due to the lack of a rich multipath propagation environment. Additionally, many of these deployments have limited range between adjacent mesh nodes due to physical obstructions resulting in the attenuation of signal levels.
Radios and systems with MIMO capabilities intended for use in both near line of sight (NLOS) and line of sight (LOS) environments are disclosed in U.S. patent application Ser. No. 13/212,036, now U.S. Pat. No. 8,238,318, and Ser. No. 13/536,927, both of which are incorporated herein by reference, and are referred to herein by the term “Intelligent Backhaul Radio” (IBR).
FIGS. 3A and 3B illustrate exemplary embodiments of the disclosed IBRs. In FIGS. 3A and 3B, the IBRs include interfaces 304A, interface bridge 308A, MAC 312A, modem 324A, channel MUX 328A, RF 332A, which includes Tx1 . . . TxM 336A and Rx1 . . . RxN 340A, IBR Antenna Array 348A (includes multiple antennas 352A), a Radio Link Controller (RLC) 356A and a Radio Resource Controller (RRC) 360A. The IBR may optionally include an “Intelligent Backhaul Management System” (or “IBMS”) agent 370B as shown in FIG. 3B. It will be appreciated that the components and elements of the IBRs may vary from that illustrated in FIGS. 3A and 3B.
Embodiments of such intelligent backhaul radios, as disclosed in the foregoing references, include one or more demodulator cores within modem 324A, wherein each demodulator core demodulates one or more receive symbol streams to produce a respective receive data interface stream; a plurality of receive RF chains 340A within IBR RF 332A to convert from a plurality of receive RF signals from IBR Antenna Array 348A, to a plurality of respective receive chain output signals; a frequency selective receive path channel multiplexer within IBR Channel multiplexer 328A, interposed between the one or more demodulator cores and the plurality of receive RF chains, to produce the one or more receive symbol streams provided to the one or more demodulator cores from the plurality of receive chain output signals; an IBR Antenna Array (348A) including: a plurality of directive gain antenna elements 352A; and one or more selectable RF connections that selectively couple certain of the plurality of directive gain antenna elements to certain of the plurality of receive RF chains, wherein the number of directive gain antenna elements that can be selectively coupled to receive RF chains exceeds the number of receive RF chains that can accept receive RF signals from the one or more selectable RF connections; and a radio resource controller, wherein the radio resource controller sets or causes to be set the specific selective couplings between the certain of the plurality of directive gain antenna elements and the certain of the plurality of receive RF chains.
The intelligent backhaul radio may further include one or more modulator cores within IBR Modem 324A, wherein each modulator core modulates a respective transmit data interface stream to produce one or more transmit symbol streams; a plurality of transmit RF chains 336A within IBR RF 332A, to convert from a plurality of transmit chain input signals to a plurality of respective transmit RF signals; a transmit path channel multiplexer within IBR Channel MUX 328A, interposed between the one or more modulator cores and the plurality of transmit RF chains, to produce the plurality of transmit chain input signals provided to the plurality of transmit RF chains from the one or more transmit symbol streams; and, wherein the IBR Antenna Array 348A further includes a plurality of RF connections to couple at least certain of the plurality of directive gain antenna elements to the plurality of transmit RF chains.
The primary responsibility of the RLC 356A in exemplary intelligent backhaul radios is to set or cause to be set the current transmit “Modulation and Coding Scheme” (or “MCS”) and output power for each active link. For links that carry multiple transmit streams and use multiple transmit chains and/or transmit antennas, the MCS and/or output power may be controlled separately for each transmit stream, chain, or antenna. In certain embodiments, the RLC operates based on feedback from the target receiver for a particular transmit stream, chain and/or antenna within a particular intelligent backhaul radio.
The intelligent backhaul radio may further include an intelligent backhaul management system agent 370B that sets or causes to be set certain policies relevant to the radio resource controller, wherein the intelligent backhaul management system agent exchanges information with other intelligent backhaul management system agents within other intelligent backhaul radios or with one or more intelligent backhaul management system servers.
FIG. 3C illustrates an exemplary embodiment of an IBR Antenna Array 348A. FIG. 3C illustrates an antenna array having Q directive gain antennas 352A (i.e., where the number of antennas is greater than 1). In FIG. 3C, the IBR Antenna Array 348A includes an IBR RF Switch Fabric 312C, RF interconnections 304C, a set of Front-ends 308C and the directive gain antennas 352C. The RF interconnections 304C can be, for example, circuit board traces and/or coaxial cables. The RF interconnections 304C connect the IBR RF Switch Fabric 312C and the set of Front-ends 308C. Each Front-end 308C is associated with an individual directive gain antenna 352A, numbered consecutively from 1 to Q.
FIG. 3D illustrates an exemplary embodiment of the Front-end circuit 308C of the IBR Antenna Array 348A of FIG. 3C for the case of TDD operation, and FIG. 3E illustrates an exemplary embodiment of the Front-end circuit 308C of the IBR Antenna Array 348A of FIG. 3C for the case of FDD operation. The Front-end circuit 308C of FIG. 3E includes a transmit power amplifier PA 304D, a receive low noise amplifier LNA 308D, SPDT switch 312D and band-select filter 316D. The Front-end circuit 308C of FIG. 3E includes a transmit power amplifier PA 304E, receive low noise amplifier LNA 308E, and duplexer filter 312E. These components of the Front-end circuit are substantially conventional components available in different form factors and performance capabilities from multiple commercial vendors.
As shown in FIGS. 3D and 3E, each Front-end 308E also includes an “Enable” input 320D, 320E that causes substantially all active circuitry to power-down. Power-down techniques are well known. Power-down is advantageous for IBRs in which not all of the antennas are utilized at all times. It will be appreciated that alternative embodiments of the IBR Antenna Array may not utilize the “Enable” input 320D, 320E or power-down feature. Furthermore, for embodiments with antenna arrays where some antenna elements are used only for transmit or only for receive, then certain Front-ends (not shown) may include only the transmit or only the receive paths of FIGS. 3D and 3E as appropriate.
As described above, each Front-end (FE-q) corresponds to a particular directive gain antenna 352A. Each antenna 352A has a directivity gain Gq. For IBRs intended for fixed location street-level deployment with obstructed LOS between IBRs, whether in PTP or PMP configurations, each directive gain antenna 352A may use only moderate directivity compared to antennas in conventional PTP systems at a comparable RF transmission frequency.
In the exemplary IBR Antenna Array 348A illustrated in FIGS. 3A, 3B and 3C, the total number of individual antenna elements 352A, Q, is greater than or equal to the larger of the number of RF transmit chains 336A, M, and the number of RF receive chains 340A, N. In some embodiments, some or all of the antennas 352A may be split into pairs of polarization diverse antenna elements realized by either two separate feeds to a nominally single radiating element or by a pair of separate orthogonally oriented radiating elements. Such cross polarization antenna pairs enable either increased channel efficiency or enhanced signal diversity as described for the conventional PTP radio. The cross-polarization antenna pairs as well as any non-polarized antennas are also spatially diverse with respect to each other. Additionally, the individual antenna elements may also be oriented in different directions to provide further channel propagation path diversity.
Additional embodiments supporting MIMO technology in specific embodiments include the use so-called zero division duplexed (ZDD) intelligent backhaul radios (ZDD-IBR), as disclosed in U.S. patent application Ser. No. 13/609,156, which is additionally incorporated herein by reference.
Embodiments of the ZDD systems provide for the operation of a IBR wherein the ZDD-IBR transmitter and receiver frequencies are close in frequency to each other so as to make the use of frequency division duplexing, as known in the art, impractical. Arrangements of ZDD operation disclosed in the foregoing referenced application include so-called “co-channel” embodiments wherein the transmit frequency channels in use by a ZDD-IBR, and the receive frequencies are partially or entirely overlapped in the frequency spectrum. Additionally disclosed embodiments of ZDD-IBRs include so-called “co-band” ZDD operation wherein the channels of operation of the ZDD-IBR are not directly overlapped with the ZDD-IBR receive channels of operation, but are close enough to each other so as to limit the performance the system. For example, at specific receiver and transmitter frequency channel separation, the frequency selectivity of the channel selection filters in an IBR transmitter and receiver chains may be insufficient to isolate the receiver(s) from the transmitter signal(s) or associated noise and distortion, resulting in significant de-sensitization of the IBR's receiver(s) performance at specific desired transmit power levels, with out the use of disclosed ZDD techniques. Embodiments of the disclosed ZDD-IBRs include the use of radio frequency, intermediate frequency and base band cancelation of reference transmitter and interference signals from the ZDD-IBR receivers in a MIMO configuration. Such disclosed ZDD techniques utilize the estimation of the channels from the plurality of IBR transmitters to the plurality of IBR receivers of the same intelligent backhaul radio, and the adaptive filtering of the reference signals based upon the channel estimates so as to allow the cancelation the transmitter signals from the receivers utilizing such estimated cancelation signals. Such ZDD techniques allow for increased isolation between the desired receive signals and the ZDD-IBR's transmitters in various embodiments including MIMO configurations.
Referring now to FIG. 4A the MIMO channel matrix is depicted. Transceiver MIMO Station 405 is in communication with MIMO Station 410 utilizing MIMO channel matrix (Eq.4-1) of FIG. 4B between the 2 stations of FIG. 4A. In an example of a two-by-two MIMO system, two spatial streams are utilized between the two MIMO stations. The channel propagation matrix of Eq.4-1 is of order M by N comprised of M rows and N columns. A particular element of the channel propagation matrix, hmn, represents the frequency response of the wireless channel from the nth transmitter to the mth receiver. Therefore each element of the channel propagation matrix H is comprised of an individual complex number, if the channel is “frequency flat,” or a complex function of frequency, if the channel is “frequency selective,” which represents the amplitude and phase of the propagation channel between one transmitter and one receiver of MIMO Stations 405 and 410. Often, the channel propagation matrix and the individual propagation coefficients are frequency selective, meaning that the complex value of the coefficients vary as a function of frequency as mentioned. In a rich, multipath scattering environment, as depicted in FIG. 4C, in which sufficient signal strength reaches an intended receiver but is scattered amongst the various structures between a particular MIMO transmitter and MIMO receiver, the spatial distribution of the arriving signals is referred to as a rich multipath environment in which there is a significant angular scattering among the receiving signals at the intended receiver.
In order to separate the MIMO streams received at an intended receiver, such as MIMO Station 410 or MIMO Station 405, the channel propagation matrix H must be determined, as known in the art. The process of determining the channel propagation matrix is often performed utilizing pilot channels, preambles, and/or symbols or other known reference information. Examples of prior art systems utilizing such techniques include IEEE 802.11n, LTE, or HSPA, as well as various embodiments of intelligent backhaul radios described in U.S. Pat. No. 8,238,818 and U.S. patent application Ser. Nos. 13/536,927 and 13/609,156, which are hereby incorporated by reference in their entireties.
In order for MIMO systems (including the foregoing mentioned MIMO systems) to support a plurality of spatial MIMO streams, the order of the propagation matrix (referenced as Eq. 4-1) must exceed the desired number of streams. While this condition is necessary, it is not sufficient. The rank of the matrix must also exceed the number of desired spatial streams. The rank of a matrix is the maximum number of linearly independent column vectors of the propagation matrix. Such terminology is known in the art with respect to linear algebra. The number of supportable MIMO streams must be less than or equal to the rank of the channel propagation matrix. When the propagation coefficients from multiple transmitters of a MIMO station to a plurality of intended receive antennas are correlated, the number of linearly independent column vectors of the channel propagation matrix H is reduced and consequently the system supports fewer MIMO streams. Such a condition often occurs in environments where a small angular spread at the desired intended receiver is present, such as is the case with a line-of-sight environment where the two MIMO stations are a significant distance apart, such that the angular resolution of the receiving antennas at MIMO Station 410 is insufficient to resolve and separate the signals transmitted from the plurality of transmitters at MIMO Station 405. Such a condition is referred to as an ill-conditioned channel matrix for the desired number of streams in the MIMO system, due to the rank of the channel propagation matrix (i.e. the number of linearly independent column vectors) being less than the desired number of MIMO streams between the two MIMO stations. The reason that the rank of the channel propagation matrix is required to be greater than or equal to the desired number of MIMO streams is related to how the individual streams are separated from one another at the intended receiving MIMO station. As is known in the art, the MIMO performance is quite sensitive to the invertability of the channel propagation matrix. Such invertability, as previously mentioned, may be compromised by the receiving antenna correlation, which may be caused by close antenna spacing or small angular spread at the intended MIMO receiver. The line-of-sight condition between two MIMO stations may result in such a small angular spread between the MIMO receivers, resulting in the channel matrix being noninvertible or degenerate. Multipath fading, which often results from large angular spreads amongst individual propagation proponents between two antennas, enriches the condition of the channel propagation matrix, making the individual column vectors linearly independent and allowing the channel propagation matrix to be invertible. The inversion of the channel propagation matrix results in weights (vectors), which are utilized with the desired receive signals to separate the linear combination of transmitted streams into individual orthogonal streams, allowing for proper reception of each individual stream from spatially multiplexed composite information streams. In a line-of-sight environment, all of the column vectors of the channel propagation matrix H may be highly correlated, resulting in a matrix rank of 1 or very close to 1. Such a matrix is noninvertible and ill-conditioned, resulting in the inability to support spatial multiplexing and additional streams (other than by the use of polarization multiplexing, which provides for only 2 streams as discussed).
FIG. 4C illustrates an exemplary deployment of intelligent backhaul radios (IBRs). As shown in FIG. 4C, the IBRs 400C are deployable at street level with obstructions such as trees 404C, hills 408C, buildings 412C, etc. between them. Embodiments of intelligent backhaul radios (IBRs) are discussed in U.S. Pat. No. 8,238,318, and co-pending U.S. patent application Ser. No. 13/536,927, the entities of which are hereby incorporated by reference. The IBRs 400C are also deployable in configurations that include point-to-multipoint (PMP), as shown in FIG. 4C, as well as point-to-point (PTP). In other words, each IBR 400C may communicate with more than one other IBR 400C.
For 3G, and especially for 4th Generation (4G), cellular network infrastructure is more commonly deployed using “microcells” or “picocells.” In this cellular network infrastructure, compact base stations (eNodeBs) 416C are situated outdoors at street level. When such eNodeBs 416C are unable to connect locally to optical fiber or a copper wireline of sufficient data bandwidth, then a wireless connection to a fiber “point of presence” (POP) requires obstructed LOS capabilities, as described herein.
For example, as shown in FIG. 4C, the IBRs 400C include an Aggregation End IBR (AE-IBR) and Remote End IBRs (RE-IBRs). The eNodeB 416C of the AE-IBR is typically connected locally to the core network via a fiber POP 420C. The RE-IBRs and their associated eNodeBs 416C are typically not connected to the core network via a wireline connection; instead, the RE-IBRs are wirelessly connected to the core network via the AE-IBR. As shown in FIG. 4C, the wireless connection between the IBRs include obstructions (i.e., there may be an obstructed LOS connection between the RE-IBRs and the AE-IBR). Note that the Tall Building 412C substantially impedes the signal transmitted from RE-IBR 400C to AR-IBR 400C. Additionally, in at least one example scenario, the tree (404C) provides unacceptable signal attenuation between an RE-IBR 400C and the AE-IBR 400C.